1. Field of the Invention
The present invention relates to a light emitting diode (LED) lighting driver. More particularly, the present invention relates to an LED lighting driver including a feedback control loop based on the mean output power of the LED sections.
2. Description of the Related Art
Linear light-emitting diode (LED) lighting drivers have been proposed in recent years. They can be directly connected to utility alternating current (AC) voltage and power up an array of LED devices without the need for bulky and expensive switching-mode LED drivers.
However, recent attempts to implement the conventional linear driver into actual LED lighting products have encountered a major power regulation issue. Namely, a linear LED lighting driver supplies insufficiency illumination power when the utility AC voltage is lower, whereas it supplies excessive illumination power when the utility AC voltage is higher.
This issue arises mainly due to there are various utility AC voltages around the world. The utility AC voltage in Japan is 100Vrms. Most Latin American countries have utility AC voltage of 110Vrms, 120Vrms, or 127Vrms. USA, Canada, and some other countries use 120Vrms. China, India, Australia, and most European countries have utility AC voltage ranging from 220Vrms to 240Vrms.
Unfortunately, a conventional linear LED lighting rated at 500 lumens in 120Vrms utility AC voltage countries, such as USA and Canada, will only provide about 370 lumens in 100Vrms utility AC voltage countries, such as Japan.
The following is a discussion of circuit structures of conventional linear LED lighting drivers and why they have poor output power regulation against input AC voltage variation.
FIG. 1 is a schematic diagram showing the basic structure of a conventional linear LED lighting driver. As shown in FIG. 1, three LED sections (LD1, LD2, and LD3) are connected in series. The voltage source 101 represents the utility AC voltage input. Bridge rectifier 102 rectifies the utility AC voltage input and outputs the rectified sinusoidal voltage input, VIN 103. The anode of LD1 is connected to the rectified sinusoidal voltage input, VIN 103. The anode of a down-stream LED section is connected to the cathode of its up-stream LED section. Please notice that each LED section may include multiple LED devices configured in various series and/or parallel connection.
LED section LD1 has a forward voltage of VF1; LED section LD2 has a forward voltage of VF2; LED section LD3 has a forward voltage of VF3. In this example, VF1=70V; VF2=35V; and VF3=35V.
The linear LED lighting driver in FIG. 1 includes the same number of shunt regulators (SR1, SR2, and SR3). Each shunt regulator SR(n) includes a high-voltage metal-oxide-semiconductor field-effect transistor (MOSFET) (Q1, Q2, Q3), a current-sense resistor (R1, R2, R3), an operational amplifier (OPA1, OPA2, OPA3), and a comparator (CMP1, CMP2, CMP3), wherein n is a positive integer. Notice that CMP1 is not necessary in this example but is shown for the purpose of general discussion.
The cathode of each LED section LD1-LD3 is coupled to the rectified sinusoidal voltage power ground 104, via shunt regulator SR(n).
Each operational amplifier OPA(n) receives a pre-determined current reference value, REF(n)=REF1*r(n) and regulates the shunt current, ISH(n), according to the current reference value REF(n).
Shunt regulator SR1 is activated when VIN 103 rises above VF1 or 70V. When VIN 103 further rises above [VF1+VF2], or 105V, shunt regulator SR2 starts to conduct current. When the voltage level due to the shunt current ISH2 flowing through R2 rises above a preset value VTH, comparator CMP2 outputs a logical high signal, which turns off shunt regulator SR1. Then if VIN 103 further rises above [VF1+VF2+VF3], or 140V, shunt regulator SR3 starts to conduct current. When the voltage level across R3 rises above the value VTH, comparator CMP3 issues a logical high signal, which turns off shunt regulators SR1 and SR2.
During the time when shunt regulator SR(n) is activated, if ISH(n)*R(n)<REF(n), OPA(n) will bring higher the gate voltage of MOSFET Q(n), allowing more current to flow through. On the other hand, when ISH(n)*R(n)>REF(n), OPA(n) will bring lower the gate voltage of MOSFET Q(n), allowing less current to flow through. By the high gain of OPA (n), typically higher than 60 dB, ISH(n) is controlled precisely at a level of ISH(n)=REF(n)/R(n), as long as there is sufficient headroom voltage across the shunt regulator SR(n).
On the other hand, comparator CMP(n) receives a threshold reference value VTH. If shunt current ISH(n) rises above VTH/R(n), comparator CMP(n) turns off all up-stream shunt regulators, SR1 to SR(n−1). If shunt current ISH(n) drops below VTH/R(n), comparator CMP(n) re-activates its immediate up-stream shunt regulator SR(n−1). Typically, VTH value is set to 20% to 50% of REF1.
Notice that when shunt regulator SR(n) is in a steady-state operation, wherein its shunt current ISH(n) is higher than VTH/R(n), all other shunt regulators are in OFF state. Since comparator CMP(n) issues a logical high signal when shunt regulator SR(n) is in a steady-state operation, the output signal of comparator CMP(n) can be used to indicate the conduction time of shunt regulator SR(n).
A high-voltage (HV) linear regulator 105 consists of a high-voltage MOSFET Q9, a pull-up resistor R9, a zener reference diode D9, and a filter capacitor C9. HV linear regulator 105 is coupled to the rectified sinusoidal voltage, VIN 103, to supply a low voltage source VCC to power the shunt regulators SR(n).
The following is a discussion regarding the square current waveform control mechanism of the conventional linear LED lighting driver in FIG. 1. Referring to FIG. 2, in this case, all 3 shunt regulators regulate at the same current level. REF1=REF2=REF3. In another word, the shunt current ratio r2=ISH2/ISH1=1.0; r3=ISH3/ISH1=1.0. As shown in FIG. 2, the LED current is maintained at 60 mA from T1 to T6.
For simplicity of discussion, assume again VF1=70V, VF2=VF3=35V. The regulation level of each shunt regulator is set up to be 60 mA equally. In FIG. 2, curve 201 represents the rectified sinusoidal voltage waveform of VIN when the AC line voltage is 100Vrms. Curve 202 represents the voltage across the active LED sections. It has discrete steps of 70V, 105V, and 140V.
From T0 to T1, when VIN 201 is less than 70V, none of the LED sections is in conduction. At T1, VIN 201 rises above 70V, LED section LD1 starts to conduct. Between T1 and T2, voltage across LD1 remains at 70V. Shunt regulator SR1 keeps ISH1 regulated at 60 mA level, with Q1 absorbing the differential voltage of VIN−70V.
At time T2, VIN 201 rises above 105V, LD2 starts to conduct. ISH2 rises quickly. Once ISH2*R2 rises above VTH level, comparator CMP2 turns off shunt regulator SR1. Between T2 and T5, the combined forward voltage across LED sections LD1 and LD2 remains 105V. Shunt regulator SR2 keeps ISH2 regulated at 60 mA level, with Q2 absorbing the differential voltage of VIN−105V.
However, since the peak voltage of VIN 201 only reaches 139.4V (=100V*1.414−2V), LED section LD3 never conducts. LED section LD2 is in conduction from T2 until T5, when VIN 201 drops below 105V again. At T5, ISH2*R2 drops below VTH level. The output of comparator CMP2 goes low, which re-activates shunt regulator SR1. At T6, VIN 201 drops below 70V, and ISH1 drops to zero.
On the other hand, when the AC line voltage is at 120Vrms, the rectified sinusoidal voltage VIN is shown in FIG. 2 as curve 204, whereas curve 205 represents the voltage across the active LED sections at different time points.
The circuit operation in the case of 120Vrms is similar to the case of 100Vrms, except LED section LD3 now conducts from T3 to T4, when VIN 204 rises above 140V. The peak voltage of a 120Vrms line voltage is near 166V. The conduction time of shunt regulator SR3 is about 3.0 ms.
In the case of VIN=100Vrms, 60 Hz, it is calculated that T1=1.412 ms, T2=2.295 ms, T5=6.038 ms, and T6=6.921 ms. The average power of the LED lighting over a half line cycle can be determined as follows,
The output power Pout=60 mA*[70V*(T12+T56)+105V*(T25)]/8.333 ms=3.720 W, wherein T12=T2−T1; T56=T6−T5; T25=T5−T2. Notice that the LED current conduction time in a half line cycle is T6−T1=6.921−1.412=5.509 ms.
In the case of VIN=120Vrms, 60 Hz, it is found that T1=1.155 ms, T2=1.817 ms, T3=2.663 ms, T4=5.670 ms, T5=6.516 ms, and T6=7.178 ms.
The LED lighting output power, Pout, now includes the contribution from LD3.
In the case of VIN=120Vrms, 60 Hz, the output power Pout=60 mA*[70V*(T12+T56)+105V*(T23+T45)+140V*(T34)]/8.333 ms=4.977 W, wherein T12=T2−T1; T23=T3−T2; T34=T4−T3; T45=T5−T4; T56=T6−T5. Please notice that the conduction time in a half line cycle is now 7.178−1.155=6.023 ms.
Assume the LED devices used in the FIG. 1 circuit has efficacy of 100 lumens per watt, in the case of 100Vrms line voltage, the light output is 372 lumens; whereas in the case of 120Vrms line voltage, it has increased to 498 lumens, or 33.8% more lumens. In comparison, the conduction time has increased from 5.509 ms to 6.023 ms, or only about 9.33% more. This disparity of LED output power is a major drawback of conventional linear LED lighting drivers.
In a location where the utility AC voltage is 127Vrms, the light output will increase to 522 lumens. That is 40.3% more lumens than the 100Vrms case. Furthermore, it is calculated that the actual input power, Pin, is increased to 6.16 W. This excessive power dissipation will make the LED sections to heat up substantially. In general, LED sections running at elevated temperature will degrade their reliability and usable life.
The following is a discussion regarding the stacked current waveform control mechanism of the conventional linear LED lighting driver in FIG. 1. Please refer to FIG. 3.
One way to improve efficiency and to make current waveform more similar to the sinusoidal waveform of the rectified sinusoidal voltage is to apply a staircase or a stacked current waveform. In this case, the shunt current ratios r2 and r3 are chosen with the following relationship: r3>r2>1.0.
As an example, the shunt regulators in the FIG. 1 circuit can have different current reference levels. For example, ISH1=60 mA, ISH2=72 mA, ISH3=84 mA. This is equivalent to setting r2=1.2, and r3=1.4. In FIG. 3, curve 301 represents the rectified sinusoidal voltage waveform of VIN when the AC line voltage is 100Vrms. Curve 302 represents the voltage across the active LED sections. Curve 303 shows the stacked current waveform at VIN=100Vrms. The LED output power is calculated as 3.953 W.
At VIN=120Vrms, the rectified sinusoidal voltage waveform of VIN is shown as curve 304 and the voltage across the active LED sections is shown as curve 305. The stacked current waveform is shown as curve 306, which has an additional 84 mA step for ISH3. The LED output power is increased to 5.423 W, or 41.1% more than the 100Vrms case.
The line regulation of the stacked current waveform is in general slightly worse than that of the square current waveform. This is expected since the power contribution from LED section LD3 is even more pronounced.
In summary, conventional LED lighting drivers have some limitations. The first limitation is poor output power regulation against AC line voltage variation. The second limitation is that LED current will drop to zero when the rectified sinusoidal voltage VIN drops below the forward voltage of the first LED section during a half line cycle.
As explained previously, the AC line voltage value greatly affects the LED output power. In fact, the average LED output power over a half line cycle can be properly described by the following equation,[VF1*ISH1*(T12+T56)+VFS2*ISH2*(T23+T45)+VFS3*ISH3*(T34)]/8.33 ms  Eq. (1)
where T12=T2−T1; T23=T3−T2; T34=T4−T3; T45=T5−T4; T56=T6−T5. VFS(n) is the combined forward voltage of LED sections LD1 through LD(n). For example, VFS2=VF1+VF2; VFS3=VF1+VF2+VF3.
Also, T1=SIN−1 (VF1/Vpk)*8.33 ms; T2=SIN−1 (VFS2/Vpk)*8.333 ms; T3=SIN−1 (VFS3/Vpk)*8.333 ms; where Vpk is the peak value of the rectified sinusoidal voltage VIN.